Method and system for a simplified user group selection scheme with finite-rate channel state information feedback for FDD multiuser MIMO downlink transmission

ABSTRACT

Certain embodiments of the invention may be found in a method and system for a simplified user group selection scheme with finite-rate channel state information feedback for frequency division duplex (FDD) multiuser multiple-input-multiple-output (MIMO) downlink transmission. Aspects of the method may include selecting, from a plurality of users, a first user having a channel gain that is greater than a channel gain corresponding to a remaining portion of the plurality of users. At least a second user may be selected from a portion of the remaining portion of the plurality of users. The selected at least the second user has a channel direction that may be approximately orthogonal to that of the first user. The channel gain may be defined by a quantized channel gain. A quantized channel direction of the selected first user may be determined for the selecting the at least the second user.

CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE

This application makes reference to:

U.S. application Ser. No. 11/232,266 filed Sep. 21, 2005;

U.S. application Ser. No. 11/231,501 filed Sep. 21, 2005;

U.S. application Ser. No. 11/231,699 filed Sep. 21, 2005;

U.S. application Ser. No. 11/231,586 filed Sep. 21, 2005;

U.S. application Ser. No. 11/232,369 filed Sep. 21, 2005;

U.S. application Ser. No. 11/231,701 filed Sep. 21, 2005;

U.S. application Ser. No. 11/232,362 filed Sep. 21, 2005;

U.S. application Ser. No. 11/231,557 filed Sep. 21, 2005; and

U.S. application Ser. No. 11/231,416 filed Sep. 21, 2005.

Each of the above stated applications is hereby incorporated herein by reference in its entirety.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to processing of signals in communication systems. More specifically, certain embodiments of the invention relate to a method and system for a simplified user group selection scheme with finite-rate channel state information feedback for frequency division duplex (FDD) multiuser multiple-input-multiple-output (MIMO) downlink transmission.

BACKGROUND OF THE INVENTION

Mobile communications have changed the way people communicate and mobile phones have been transformed from a luxury item to an essential part of every day life. The use of mobile phones is today dictated by social situations, rather than hampered by location or technology. While voice connections fulfill the basic need to communicate, and mobile voice connections continue to filter even further into the fabric of every day life, the mobile Internet is the next step in the mobile communication revolution. The mobile Internet is poised to become a common source of everyday information, and easy, versatile mobile access to this data will be taken for granted.

Third generation (3G) cellular networks have been specifically designed to fulfill these future demands of the mobile Internet. As these services grow in popularity and usage, factors such as cost efficient optimization of network capacity and quality of service (QoS) will become even more essential to cellular operators than it is today. These factors may be achieved with careful network planning and operation, improvements in transmission methods, and advances in receiver techniques. To this end, carriers need technologies that will allow them to increase downlink throughput and, in turn, offer advanced QoS capabilities and speeds that rival those delivered by cable modem and/or DSL service providers.

In order to meet these demands, communication systems using multiple antennas at both the transmitter and the receiver have recently received increased attention due to their promise of providing significant capacity increase in a wireless fading environment. These multi-antenna configurations, also known as smart antenna techniques, may be utilized to mitigate the negative effects of multipath and/or signal interference on signal reception. It is anticipated that smart antenna techniques may be increasingly utilized both in connection with the deployment of base station infrastructure and mobile subscriber units in cellular systems to address the increasing capacity demands being placed on those systems. These demands arise, in part, from a shift underway from current voice-based services to next-generation wireless multimedia services that provide voice, video, and data communication.

The utilization of multiple transmit and/or receive antennas is designed to introduce a diversity gain and to raise the degrees of freedom to suppress interference generated within the signal reception process. Diversity gains improve system performance by increasing received signal-to-noise ratio and stabilizing the transmission link. On the other hand, more degrees of freedom allow multiple simultaneous transmissions by providing more robustness against signal interference, and/or by permitting greater frequency reuse for higher capacity. In communication systems that incorporate multi-antenna receivers, a set of M receive antennas may be utilized to null the effect of (M-1) interferers, for example. Accordingly, N signals may be simultaneously transmitted in the same bandwidth using N transmit antennas, with the transmitted signal then being separated into N respective signals by way of a set of N antennas deployed at the receiver. Systems that utilize multiple transmit and receive antennas may be referred to as multiple-input multiple-output (MIMO) systems. One attractive aspect of multi-antenna systems, in particular MIMO systems, is the significant increase in system capacity that may be achieved by utilizing these transmission configurations. For a fixed overall transmitted power, the capacity offered by a MIMO configuration may scale with the increased signal-to-noise ratio (SNR). For example, in the case of fading multipath channels, a MIMO configuration may increase system capacity by nearly M additional bits/cycle for each 3-dB increase in SNR.

The widespread deployment of multi-antenna systems in wireless communications has been limited by the increased cost that results from increased size, complexity, and power consumption. This poses problems for wireless system designs and applications. As a result, some initial work on multiple antenna systems may be focused on systems that support single user point-to-point links. However, the use of multi-antenna techniques for a multiuser environment to improve total throughput remains a challenge.

Communication systems using multiple antennas at both the transmitter and the receiver have recently received increased attention due to their promise of providing significant capacity increase in a wireless fading environment. However, most of the initial work on multiple antenna systems was focused on systems that support single user point-to-point links. More recently, various efforts have focused on utilizing multi-antenna techniques to a multiuser environment to improve total throughput.

The performance of conventional multiple antenna systems may depend on the availability of channel state information (CSI) at the transmitter (CSIT) and at the receiver (CSIR). During design and analysis stage, most of the conventional multiple-input-multiple-output (MIMO) communication systems may utilize an assumption that CSIT is complete or that there is no CSIT information available. However, in most instances, such assumptions may be impractical as only quantized CSI may be available at the transmitter. Communication systems with multiple users, such as a CDMA-based communication system or communication systems within a wireless LAN environment, may utilize user group selection algorithms and CSI feedback to achieve array gain and transmit diversity. Furthermore, multiuser diversity of the downlink channel may also be achieved by using CSI information, which may be available at the transmitter. However, most conventional user group selection algorithms, such as the optimal brute-force user group selection algorithm, may utilize complex computations to determine the CSI and may require feedback of the entire CSI from the receiving stations to the transmitting base station. Such computation complexity and full CSI feedback may increase implementation costs and reduce processing time and overall efficiency of the communication system.

Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of skill in the art, through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings.

BRIEF SUMMARY OF THE INVENTION

A system and/or method is provided for a simplified user group selection scheme with finite-rate channel state information feedback for frequency division duplex (FDD) multiuser multiple-input-multiple-output (MIMO) downlink transmission, substantially as shown in and/or described in connection with at least one of the figures, as set forth more completely in the claims.

These and other advantages, aspects and novel features of the present invention, as well as details of an illustrated embodiment thereof, will be more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1A is a diagram illustrating exemplary signal transmission from base station to receivers within a finite-rate feedback multiuser communication environment, in accordance with an embodiment of the invention.

FIG. 1B is a top-level block diagram illustrating an exemplary multiuser multiple-input-multiple-output (MIMO) downlink transmission system with finite-rate channel state information feedback, in accordance with an embodiment of the invention.

FIG. 2A is a high level flow diagram illustrating exemplary steps for user group selection with finite-rate channel state information feedback, in accordance with an embodiment of the invention.

FIG. 2B is a flow diagram illustrating exemplary steps for user group selection with finite-rate channel state information feedback, in accordance with an embodiment of the invention.

FIG. 3 is a graph that illustrates exemplary downlink transmission schemes in terms of sum rate, in accordance with an embodiment of the present invention.

FIG. 4 is a graph that illustrates exemplary downlink transmission schemes in terms of bit error rate, in accordance with an embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method and system for a simplified user group selection scheme with finite-rate channel state information feedback for frequency division duplex (FDD) multiuser multiple-input-multiple-output (MIMO) downlink transmission. Aspects of the method may comprise selecting, from a plurality of users, a first user having a channel gain that is greater than a channel gain corresponding to a remaining portion of the plurality of users. At least a second user may be selected from a portion of the remaining portion of the plurality of users. The selected at least the second user has a channel direction that may be approximately orthogonal to that of the first user. The channel gain may be defined by a quantized channel gain. A quantized channel direction of the selected first user may be determined for the selecting the at least the second user. In this regard, a simplified user group selection scheme in accordance with an embodiment of the invention may significantly reduce the computational complexity and the total amount of CSI feedback information communicated from a receiver station to a base station transmitter. As used herein, the term “user” may be interpreted to refer to a mobile device.

FIG. 1A is a diagram illustrating an exemplary signal transmission from base station to receivers within a finite-rate feedback multiuser communication environment, in accordance with an embodiment of the invention. Referring to FIG. 1A, there is shown a communication system 100 a comprising a base station 102 a, a first user (user 1) 104 a, and a second user (user 2) 106 a. The base station 102 a may comprise antennas 116 a, . . . , 122 a. The first user 104 a and the second user 106 a may each have a single antenna. In this instance, the base station 102 a may transmit signals 112 a and 114 a, which are intended to be received by the first user 104 a and by the second user 106 a, respectively.

After the first user 104 a and the second user 106 a process signals 112 a and 114 a, respectively, the first user 104 a and the second user 106 a may communicate quantized channel state information ĥ₁ 108 a and ĥ₂ 110 a. The quantized channel state information ĥ₁ 108 a and ĥ₂ 110 a may comprise channel gain information and channel direction information for the first user 104 a and the second user 106 a, respectively. The base station 102 a may utilize the received quantized channel state information ĥ₁ 108 a and ĥ₂ 110 a to precode transmit information for the first user 104 a and the second user 106 a. In one embodiment of the invention, the first user 104 a and the second user 106 a may communicate limited or finite-rate channel state information 108 a and 110 a to the base station 102 a via the rate constraint feedback link between the base station 102 a and the users 104 a and 106 a, in order to maximize throughput and increase processing speed and efficiency.

In another embodiment of the invention, the base station 102 a may be equipped with M antennas and there may be K users within the communication system 100 a, where each user may comprise a single antenna. In such circumstances, the signal model may be expressed as

$\begin{matrix} {{\begin{bmatrix} y_{1} \\ y_{2} \\ \vdots \\ y_{k} \end{bmatrix} = {{\begin{bmatrix} h_{1} \\ h_{2} \\ \vdots \\ h_{k} \end{bmatrix}x} + n}},} & (1) \end{matrix}$ where y_(k) (k=1, . . . , K) may be the signal received by user k, h_(k)ε

^(1×M) may be the channel vector to user k, xε

^(M×1) may be the transmitted symbol vector by the base station 102 a, and nε

^(k×1) may be the additive white Gaussian noise (AWGN) with zero mean and unit variance. The transmitted symbols may satisfy certain power constraint, for example where E[x^(H)x]≦P, where (•)^(H) may represent complex conjugate transpose.

In this exemplary analysis, each element in h_(k) may be a zero-mean circularly symmetric complex Gaussian (ZMCSCG) random variable with unit variance. Moreover, the users may experience independent fading and, therefore, the channel vectors {h_(k)}_(k=1) ^(k) may be statistically independent from each other. The channel state information (CSI), h_(k), may be known to user k, but may not be known to other users. The base station 102 a may have knowledge of the CSI for all users. In one embodiment of the invention, the communication system 100 a may be a frequency division duplex (FDD) communication system. In this regard, the base station 102 a may obtain CSI 108 a and 110 a from the first user 104 a and the second user 106 a through a rate constraint feedback link.

Employing multiple antennas at the base station in cellular multiuser communication systems may improve the downlink system capacity. This approach may be utilized with any multiuser MIMO system, such as code division multiple access 2000 (CDMA2000), wideband CDMA (WCDMA), and Wireless LAN (WLAN), for example. The capacity improvement may be attained by communicating simultaneously to multiple users through precoding at the transmitter or base station when channel state information is available. In this regard, a transmitter or base station may refer to any device or equipment which may be adapted to communicate with multiple other devices, users, and/or receivers. Moreover, a user or a receiver may refer to user equipment or device that may be adapted for communication with a base station and/or other devices. Dirty paper coding (DPC) may be utilized as a precoding scheme that may achieve the sum capacity. However, DPC may be difficult to implement due to complexity issues. There may also exist other suboptimal but relatively low complexity schemes for multiuser MIMO downlink, such as linear precoding, Tomlinson-Harashima precoding, and vector encoding, for example.

A zero-forcing (ZF) linear precoder may achieve the sum capacity when combined with infinite-order multiuser diversity, that is, when the number of users K approaches infinity. Moreover, ZF precoders may provide near-optimal performance even with a limited number of users, when K=10 for example.

The zero-forcing precoders may be a specific type of linear precoders. When the base station, for example the base station 102 a in FIG. 1A, decides to transmit to a group of users D⊂{1, . . . , K} with d=|D|≦k, a linear precoding scheme may linearly weigh the data symbols s=[s₁, . . . , s_(d)]^(T) before they are transmitted from the base station, x=FP_(s)  (2) where x is the transmitted signal vector as in equation (1), F=[f₁, . . . , f_(d)] may be the M×d linear preceding matrix with normalized columns (∥f_(k)∥=1), and P=diag{P₁, . . . , P_(d)} with

${\sum\limits_{i = 1}^{d}P_{i}} \leq P$ may be the power control matrix that may be adapted to allocate transmit power to different users. The received signal may be given by equation

$\begin{matrix} {\begin{bmatrix} y_{1} \\ y_{2} \\ \vdots \\ y_{d} \end{bmatrix} = {{\begin{bmatrix} h_{1} \\ h_{2} \\ \vdots \\ h_{d} \end{bmatrix}{FPs}} + {n.}}} & (3) \end{matrix}$

A zero-forcing precoder may use the pseudo-inverse of the overall channel matrix H_(D)=[h₁ ^(T), . . . , h_(d) ^(T)]^(T) as the weighting matrix when H_(D) has full row rank, i.e.

$\begin{matrix} {{W_{D} = {H_{D}^{\dagger} = {H_{D}^{H}\left( {H_{D}H_{D}^{H}} \right)}^{- 1}}},} & (4) \\ {{F_{D} = {W_{D}\begin{bmatrix} \frac{1}{w_{1}} & \; & \; \\ \; & ⋰ & \; \\ \; & \; & \frac{1}{w_{d}} \end{bmatrix}}},} & (5) \end{matrix}$ where {w_(i)}_(i=1) ^(d) are the column of W_(D)

By defining

$\begin{matrix} {\xi_{i}\overset{\bigtriangleup}{=}\frac{1}{w_{i}}} & (6) \end{matrix}$ and substituting equation (5) into equation (3), the received signal for each user with zero-forcing precoding may be expressed as, y _(i)=ξ_(i) P _(i) s _(i) +n _(i) , ∀iεD.  (7)

In this regard, the multiuser downlink channel may become a set of parallel channels. The maximum sum rate of the given user group D may be given by

$\begin{matrix} {{C_{D} = {\sum\limits_{i \in D}{\log\left( {1 + {\xi_{i}P_{i}}} \right)}}},} & (8) \end{matrix}$ where the optimal P_(i) may be given by the water-filling solution,

$\begin{matrix} {{P_{i} = \left( {\mu - \frac{1}{\xi_{i}}} \right)^{+}},} & (9) \end{matrix}$ with the water level μ chosen to satisfy

${\sum\limits_{i \in D}\left( {\mu - \frac{1}{\xi_{i}}} \right)^{+}} = {P.}$ The maximum achievable sum rate for a given channel realization may be obtained by searching over all the possible user groups, that is,

$\begin{matrix} {C = {\max\limits_{{D \subseteq {\{{1,\ldots\mspace{11mu},K}\}}},{{D} \leq M}}C_{D}}} & (10) \end{matrix}$

An optimal or best user group selection for ZF precoding may require searching over a plurality of

$\sum\limits_{i = 1}^{M}\begin{pmatrix} K \\ i \end{pmatrix}$ candidate user groups to find the one with the largest sum rate, which leads to a fairly high computational cost. Moreover, in an FDD system such as the communication system 100 a, the channel state information that may be needed at the transmitter 102 a to perform the optimal user group search may be obtained from the users 104 a and 106 a through a feedback link. Because the optimal search requires CSI from each user and each user's channel is a complex vector of dimension M, that is equivalent to 2M real numbers per user, heavy burden may be placed on the feedback link to obtain this information. This may be particularly cumbersome since the feedback link tends to have very limited capacity. A user group selection scheme, such as the selection scheme described below in reference to FIG. 2, may result in a simpler implementation and may require less feedback information. In this regard, such selection scheme may be utilized for finite-rate channel state information feedback within multiuser communication systems with multiple transmits antennas.

FIG. 1B is a top-level block diagram illustrating an exemplary multiuser multiple-input-multiple-output (MIMO) downlink transmission system with finite-rate channel state information feedback, in accordance with an embodiment of the invention. Referring to FIG. 1B, there is shown a communication system 100 b that may comprise a base station 102 b and a plurality of K users 150 b, . . . , 152 b. Within the communication system 100 b, the base station 102 b may comprise M transmit (TX) antennas and each of the K users 150 b, . . . , 152 b may each have a single receive (RX) antenna. In this implementation, the total number of users or receiver antennas may be equal or higher than the number of base station antennas, that is, K≧M.

The base station 102 a may comprise a plurality of channel encoders 104 b, . . . , 106 b, a user scheduler 108 b, a plurality of modulators (MOD) 110 b, . . . , 112 b, a power control block 114 b, a beamforming or linear precoding block 116 b, a processor 107 b, and a memory 109 b. Each of the plurality of users 150 b, . . . , 152 b may each comprise one of a plurality of demodulators (DEM) 118 b, . . . , 119 b, one of a plurality of channel decoders 120 b, . . . , 121 b, one of a plurality of channel estimators 122 b, . . . , 123 b, one of a plurality of channel quantizers 126 b, . . . , 127 b, and one of a plurality of feedback controllers 124 b, . . . , 125 b.

The channel encoders 104 b, . . . , 106 b may comprise suitable logic, circuitry, and/or code that may be adapted to encode input binary data b₁, . . . , b_(k) for each of the K users in the communication system 100 b. The beamforming or linear precoding block 116 b may comprise suitable logic, circuitry, and/or code that may be adapted to processes the user data symbols to separate signals intended for different users such that each user receives little or no interference from other users. With M antennas at the base station 102 b, the beamforming or linear precoding block 116 b may separate at most M different signals, that is, the base station 102 b may transmit to at most M users at a time. Therefore, for each channel realization, the base station 102 b may need to select M or less than M users among all the K users to transmit.

The user scheduler 108 b may comprise suitable logic, circuitry, and/or code that may be adapted to locate a best user group from the plurality of users 150 b, . . . , 152 b that optimizes certain performance criterion such as the sum throughput of the system, for example. In this regard, the user scheduler 108 b may be adapted to perform the steps of a simplified user group selection algorithm, for example, to find the best user group. The user scheduler 108 b may utilize knowledge of the channel state information (CSI) provided by the users 150 b, . . . , 152 b via the feedback path 142 b when determining the best user group. The user scheduler 108 b may be adapted to select a first user with the strongest channel gain and a second user with the second strongest channel gain. The user scheduler 108 b may be adapted to determine a first maximum system capacity based on the first user and a second maximum system capacity based on the second user. The user scheduler 108 b may also be adapted to select the highest of the first maximum system capacity and the second maximum system capacity as the maximum system capacity to be supported by the communication system 100 b. In this regard, for a case when M=2, the user scheduler 108 b may select the user group to comprise a pair of users associated with the maximum system capacity selected.

The modulators 110 b, . . . , 112 b may comprise suitable logic, circuitry, and/or code that may be adapted to modulate the binary data of each of the users selected by the user scheduler 108 b. In this regard, the modulation operation on the binary data may result in a plurality of complex symbols u₁, . . . , u_(M), for example. The power control block 114 b may comprise suitable logic, circuitry, and/or code that may be adapted to allocate different users with different power levels in accordance with their respective channel quality, for example, based on CSI received via the feedback path 142 b.

The user scheduler 108 b, the power control block 114 b, and/or the beamforming or linear precoding block 116 b may require knowledge of the state of the downlink channel. The processor 107 b may comprise suitable logic, circuitry, and/or code that may be adapted to process information and/or data associated with the generation of transmission signals at the base station 102 b. The processor 107 b may also be adapted to control at least a portion of the operations of the base station 102 b. The memory 109 b may comprise suitable logic, circuitry, and/or code that may be adapted to store data and/or control information that may be utilized in the operation of at least a portion of the base station 102 b.

The demodulators 118 b, . . . , 119 b in the users 150 b, . . . , 152 b may comprise suitable logic, circuitry, and/or code that may be adapted to demodulate the signals received from the base station 102 b, for example. The channel decoders 120 b, . . . , 121 b may comprise suitable logic, circuitry, and/or code that may be adapted to decode the demodulated signals from the demodulators 118 b, . . . , 119 b into received binary bit streams b′₁, . . . , b′_(k), for example.

The channel estimators 122 b, . . . , 123 b may comprise suitable logic, circuitry, and/or code that may be adapted to estimate channel state information for one or more receive channels. The channel quantizers 126 b, . . . , 127 b may comprise suitable logic, circuitry, and/or code that may be adapted to quantize channel state information estimated by the channel estimators 122 b, . . . , 123 b. The feedback controllers 124 b, . . . , 125 b may comprise suitable logic, circuitry, and/or code that may be adapted to generate channel state information output 138 b, . . . , 140 b for communication to the base station 102 b via the feedback link 142 b.

In operation, input signals b₁, . . . , b_(k) may be encoded by the channel encoders 104 b, . . . , 106 b. Based on the knowledge of the downlink channel state information 138 b, . . . , 140 b received from the users 150 b, . . . , 152 b via the feedback link 142 b, the user scheduler 108 b may select a determined group of users to transmit so as to optimize certain performance criterion, such as the sum throughput of the communication system 100 b. The binary data of each selected user may be modulated by the modulators 110 b, . . . , 112 b and may be transformed into complex symbols u₁, . . . , u_(M). The power control block 114 b may then allocate the total transmit power to different users according to their respective channel quality. The channel quality may be determined based on the downlink channel state information 138 b, . . . , 140 b received from the users 150 b, . . . , 152 b via the feedback link 142 b. The linear precoder 116 b may then process the user data symbols in such a way that each user 150 b, . . . , 152 b may receive its own signal and little or no interference from other users. After the signal is transmitted from the M base station TX antennas and after it arrives at each of the users 150 b, . . . , 152 b, it may be demodulated and decoded into received binary bit streams b′₁, . . . , b′_(k).

In a frequency division duplex (FDD) system, such as the communication system 100 b, the base station 102 b may obtain the downlink channel state information 138 b, . . . , 140 b through a finite-rate feedback link 142 b from the users 150 b, . . . , 152 b. In one embodiment of the invention, each user may estimate its own channel and may quantize the channel according to the feedback rate constraint of the communication system 100 b. In this regard, a selection algorithm may be utilized so that the feedback controller 124 b, . . . , 125 b at each user may determine a finite-rate channel state information to feed back on the request of the base station 102 b so as to optimize the user group selection by the base station 102 b. The selection algorithm which is described below with respect to FIG. 2B, may significantly reduce the communication system complexity as well as the total amount of required feedback information, thereby effectively reducing the feedback rate.

FIG. 2A is a high level flow diagram illustrating exemplary steps for user group selection with finite-rate channel state information feedback, in accordance with an embodiment of the invention. Referring to FIGS. 1B and 2A, at 202 a, the base station 102 b may select a first user from the k users 150 b, . . . , 152 b based on quantized channel gain information from all users. At 204 a, after the first user is selected, a portion of the remaining users may be selected that satisfy a determined orthogonality criteria with respect to the first user. For example, the portion of remaining users may be selected using an orthogonality threshold parameter so that the channel directions of the selected portion of remaining users may be approximately orthogonal to the channel direction of the first user. At 206 a, a second user may be selected from the selected portion of remaining users based on channel gain information. At 208 a, the first and/or the second user may be selected for transmission, based on a system capacity function. A precoding matrix may then be calculated by the base station 102 b, based on whether the first and/or the second user is selected to receive signals within the communication system 100 b.

FIG. 2B is a flow diagram illustrating exemplary steps for user group selection with finite-rate channel state information feedback, in accordance with an embodiment of the invention. Referring to FIGS. 1B and 2B, a simplified user group selection scheme may be utilized in a multiuser downlink channel with quantized CSI feedback within the communication system 100 b. For simplicity, it may be assumed that M=2. Even though the algorithm is described with respect to M=2, the present invention may not be so limited and the algorithm described herein may be applicable to other FDD communication systems with a different number of transmit antennas. As demonstrated in FIG. 2B, the user group selection algorithm may be combined with a CSI feedback scheme and may comprise the following 9 steps.

It may be assumed that users have ideal CSIR in a sense that the Multiple Input Single Output (MISO) channel impulse response h_(k)εC^(1×2) (M=2) may be known at each user. At 202, each user may quantize its own channel gain, or channel power γ_(k)=∥h_(k)∥² by a finite-rate scalar quantizer with quantization resolution B_(g) bits per channel update using the channel quantizers 126 b, . . . , 127 b. The quantized channel gain {circumflex over (γ)}_(k) or the equivalent quantization index may then be conveyed back to the base station 102 b through a rate constraint feedback link 142 b. The channel gain quantizers 126 b, . . . , 127 b may be optimized to match to the precoder 116 b used at the transmitter 102 b, such as zero-forcing precoder. One or more performance metric, such as capacity and bit error rate, as well channel statistical distributions may also be utilized in order to improve the system performance.

At 204, based upon the feedback information {circumflex over (γ)}_(κ)|_(κ=1) ^(K), the transmitter 102 b may select the user with the largest channel gain based on the following expression

$\begin{matrix} {i = {\arg\;{\max\limits_{1 \leq k \leq K}{\hat{\gamma}}_{k}}}} & (11) \end{matrix}$

Based on the selected index i, the base station may request a channel direction feedback from the i^(th) user asking for the unit norm vector v_(i)(v_(i)=h_(i)/∥h_(i)∥).

At 206, based on the request from the base station, the i^(th) user may quantize its own channel direction by a vector quantizer with inner product quantization criterion. The direction vector v_(i) may be quantized into {circumflex over (v)}_(i) with quantization resolution B_(v) bits per channel update. The quantized vector {circumflex over (v)}_(i) or the quantization index may be fed back to the base station using the same feedback link 142 b.

At 208, based on the feedback direction {circumflex over (v)}_(i), the base station 102 b may form an orthonormal beam {circumflex over (v)}_(i) ^(⊥), which may be given by the equation

$\begin{matrix} {{{\hat{v}}_{i}^{\bot} = \left\lbrack {{\hat{v}}_{i,2}^{*},{- {\hat{v}}_{i,1}^{*}}} \right\rbrack},} & (12) \end{matrix}$ where {circumflex over (v)}_(i,1) and {circumflex over (v)}_(i,2) may be the first and second elements of vector {circumflex over (v)}_(i). The base station 102 b may then broadcasts the conjugate beam {circumflex over (v)}_(i) ^(⊥)* to all the users 150 b, . . . , 152 b.

At 210, each receiver may measure its own received SNR η_(κ) with respect to the broadcast beam that the base station 102 b sends out, which may be given by the equation

$\begin{matrix} {\eta_{\kappa} = {\left\langle {h_{k},{\hat{v}}_{i}^{\bot}} \right\rangle }^{2}} & (13) \end{matrix}$

The inner product between the k^(th) user's channel directional vector v_(k) and vector {circumflex over (v)}_(i) ^(⊥) may, therefore, be given by the equation

$\begin{matrix} {{\alpha_{\kappa} = {\frac{{\left\langle {h_{k},{\hat{v}}_{i}^{\bot}} \right\rangle }^{2}}{{h_{\kappa}}^{2}} = \frac{\eta_{\kappa}}{\gamma_{\kappa}}}},} & (14) \end{matrix}$ Where α_(κ) may be an orthogonal parameter of the k^(th) user and may comprise a real number in the range [0, 1]. The value of α_(κ) may be indicative of the orthogonality of the channel directions between user j and user k. For example, α_(k)=1 may correspond to the case that h_(k) and {circumflex over (v)}_(i) may be ideally orthogonal to each other. Based on the value of α_(k), each user may form a 1 bit flag information flag_(k). The flag information flag_(k) may indicate whether the k^(th) user is orthogonal “enough” to the i^(th) user by comparing α_(k) with a fixed or preset threshold α_(th), i.e.

$\begin{matrix} {{flag}_{\kappa} = \left\{ {\begin{matrix} 1 & {\alpha_{\kappa} \geq \alpha_{th}} \\ 0 & {\alpha_{\kappa} < \alpha_{th}} \end{matrix}.} \right.} & (15) \end{matrix}$

This one bit indication information may be viewed as a flag that partitions the users into two groups: users that are semi-orthogonal to the i^(th) user, and users that are not. The flag information may be also fed back to the base station 102 b through the feedback link 142 b.

At 212, based on the feedback flag information flag_(k) and the quantized channel gain information {circumflex over (γ)}_(k) of the k^(th) user, the base station 102 b may select the second user j by selecting the strongest user in the semi-orthogonal group, i.e.

$\begin{matrix} {{j = {\arg\;{\max\limits_{\kappa \in S}{\hat{\gamma}}_{\kappa}}}},} & (16) \end{matrix}$ where

is the semi-orthogonal set defined by the following expression:

={k|flag_(κ)=1}.  (17)

In this regard, the base station 102 b has obtained two candidate users, i and j, and may select between various transmission modes. In one transmission mode, transmission to only the i^(th) user using full power may occur. In another transmission mode, transmission to both user i and user j may occur with power being equally allocated between both user i and user j. At 214, the system capacity for these transmission modes may be calculated utilizing the following expressions: C(i)≈log₂(1+ρ·{circumflex over (γ)}_(i)),  (18) C(i,j)≈log₂(1+ρ·{circumflex over (γ)}_(i)/2)+log₂(1+ρ·{circumflex over (γ)}_(j)/2),  (19) where ρ may be the system average SNR. The approximations “≈” may result from the fact that both {circumflex over (γ)}_(i) and {circumflex over (γ)}_(j) are quantized, and the two users i and j may be assumed to be perfectly orthogonal. Based on the comparison between C(i, j) and C(j), transmission mode with higher system capacity may be selected by the base station 102 b. In one embodiment of the invention, it may be assumed that the quantized channel direction {circumflex over (v)}_(j) of the second user j is orthogonal to the quantized channel direction {circumflex over (v)}_(i) of the first user i.

At 218, if C(i, j)>C(j), the transmission mode that utilizes spatial multiplexing may be utilized to communicate to both users i and j at the same time. The transmitter preceding matrix F may be formed by the following equation

$\begin{matrix} {F = {\left\lbrack {{\hat{v}}_{i}^{H}{\hat{v}}_{i}^{\bot H}} \right\rbrack/{\sqrt{2}.}}} & (20) \end{matrix}$

At 216, if C(i, j)≦C(i), a transmission mode in which the base station only communicates to the i^(th) user may be utilized. In this case, the precoding matrix may be selected according to the equation F={circumflex over (v)}_(i) ^(H).  (21)

Referring to FIG. 2B, steps 202, 206, and 210 may be achieved at the receiver side and steps 204, 208, 212, 214, 216, and 218 may be achieved at the transmitter or base station side. The approach described herein may be extended to other more sophisticated precoders that may offer better system performance such as the minimum mean square error (MMSE) precoder, the Tomlinson-Harashima preceding (THP) precoder, and/or the sphere encoding precoder, for example. When the number of users K is large enough, there may be a high probability that two users exist having close to orthogonal channel vectors. In this instance, the zero-forcing precoder may be close to an optimal precoder and most preceding techniques may have similar system performance.

A comparison in terms of both search complexity and feedback load between an optimal user group selection algorithm and the user group selection algorithm described herein may be shown in Table 1.

TABLE 1 Complexity And Feedback Comparison Brute Force User Group The Present Scheme Selection Complexity ≦2 K_(a) $\geq {\frac{K^{2}}{2}b}$ Feedback Amount K · B_(g) + K + B_(v) K · B_(g) + K · B_(v)

Table 1 may provide a summary of the comparison between the optimal user group selection algorithm and the user group selection algorithm described herein in terms of both search complexity and feedback load. The coefficient a may represent the complexity of comparing two values, and b may represent the complexity of computing water-filling solution. In this regard, b>>a. The selection algorithm described herein may only require two searches for the maximum value determination in step 202 and 212, respectively. The first search may be among K users, and the second search may be among less than K users. If a represents the complexity of comparing two positive real numbers, then the complexity of the channel state information feedback scheme disclosed herein may be given by the following expression: χ≦2Ka,  (22) as shown in Table 1. As for the optimal group search algorithm, for each of the

${\begin{pmatrix} K \\ 1 \end{pmatrix} + \begin{pmatrix} K \\ 2 \end{pmatrix}} = \frac{K\left( {K + 1} \right)}{2}$ possible groups, a water-filling solution may be calculated to obtain the sum rate for each group. If b represents the complexity of computing water-filling solution, the complexity of the optimal group selection algorithm may be represented by the expression:

$\begin{matrix} {\chi_{opt} \geq {\frac{K^{2}}{2}{b.}}} & (23) \end{matrix}$ It may be noted that a<<b, and χ<<χ_(opt).  (24)

The total amount of feedback of the channel state information feedback scheme disclosed herein may be represented by the expression: B=K·B _(g) +K+B _(v) bits,  (25) whereas that of the optimal group selection algorithm may be represented by the expression: B _(opt) =K·B _(g) +K·B _(v) bits.  (26) Therefore the required amount feedback may be significantly reduced in the new scheme described herein, B<<B_(opt).  (27)

FIG. 3 is a graph that illustrates exemplary downlink transmission schemes in terms of sum rate, in accordance with an embodiment of the present invention. Referring to FIG. 3, there are shown the results of a numerical simulation corresponding to the sum rate of a cellular system with a single base station and K=100 users. The base station may be equipped with M=2 antennas, for example, and each user may be equipped with a single antenna. The channels are flat fading Rayleigh channels. The transmit antennas at the base station may be placed sufficiently apart so as to experience independent fading. The modulation format applied in this instance may be quadrature phase-shift keying (QPSK), for example, and the orthogonality threshold α_(th), as in equation (15), may be selected to be 0.9.

Referring to FIG. 3, four results are illustrated in the graph. The signal 302 may correspond to the rate sum of a TDMA scheme with channel state information at the transmitter (CSIT). The signal 304 may correspond to the rate sum of a zero-forcing (ZF), brute-force (BF) search with ideal CSIT. In accordance with an embodiment of the invention, the signal 306 may correspond to the rate sum of a ZF, double search scheme with ideal CSIT. The signal 308 may correspond to the rate sum of a ZF, double search scheme with 3-bits allocated for channel gain feedback (Bg=3) and 6 bits allocated for channel directional feedback (Bv=6).

The TDMA scheme illustrated by the signal 302 may pick a single user in a round robin fashion and may only transmit to one user at a time using maximum ratio transmission whereas zero-forcing precoding schemes, as described by the signals 404, 406, and 408, may communicate to multiple users at a time. Therefore, the zero-forcing precoding schemes may achieve a higher sum rate than TDMA schemes. The zero-forcing precoding with the optimal brute-force user group selection, the signal 304, may exhibit the best performance at the cost of high computational complexity and high feedback load. In accordance with an embodiment of the invention, the signal 306 significantly outperforms the TDMA scheme with only very limited extra CSI feedback. It may also be noted from FIG. 3 that allocating 3 bits for channel gain feedback and 6 bits for channel direction feedback may be sufficient to approach the performance of ideal CSI at the transmitter (CSIT).

FIG. 4 is a graph that illustrates exemplary downlink transmission schemes in terms of bit error rate, in accordance with an embodiment of the present invention. Referring to FIG. 4, there are shown the results of a numerical simulation corresponding to the bit error rate (BER) of a cellular system with a single base station and K=100 users. The base station may be equipped with M=2 antennas, for example, and each user may be equipped with a single antenna. The channels may be flat fading Rayleigh channels. The transmit antennas at the base station may be spaced or separated so as to experience independent fading. The modulation format applied in this instance may be QPSK, for example.

In this instance, there are four results provided. The signal 502 may correspond to the BER of a TDMA scheme with CSIT. The signal 504 may correspond to the BER of a ZF, BF search with ideal CSIT. The signal 506 may correspond to the BER of a ZF, double search scheme with ideal CSIT. The signal 508 may correspond to the BER of a ZF, double search scheme with Bg=3 and Bv=6. The results for the BER performance between the signals 402, 404, 406, and 408 are substantially the same as the results for the signals 302, 304, 306, and 308 shown for the rate sum in FIG. 3.

In comparison to the brute-force user selection algorithm, the channel state information feedback scheme disclosed herein may significantly reduce the computational complexity. For example, the complexity may be reduced from about K²/2 to less than 2K(K>>2), where K may represent the number of users. The various embodiments of the invention may also significantly reduce the total amount of feedback information from (K·B_(g)+K·B_(v)) bits to less than (K·B_(g)+K+B_(v)) bits. Therefore, the new scheme may only require very limited CSI feedback rate, in an order of a few bits per user, for example.

Although with low rate constraint CSI feedback, the channel state information feedback scheme disclosed herein may achieve multiuser diversity as well as spacial multiplexing when there are a large amount of users. In accordance with an embodiment of the invention, this simplified user group selection scheme may significantly outperform the TDMA scheme and achieve performance in terms of capacity close to systems with ideal CSIT with minimal, about 2 bits per user, feedback rate. In one embodiment of the invention, the channel state information feedback scheme disclosed herein may be utilized with more sophisticated precoders, such as MMSE precoder, THP precoder, sphere encoding precoder, for example.

Another embodiment of the invention may provide a non-transitory machine-readable storage, having stored thereon, a computer program having at least one code section executable by a machine, thereby causing the machine to perform the steps as described above for a simplified user group selection scheme with finite-rate channel state information feedback for frequency division duplex (FDD) multiuser multiple-input-multiple-output (MIMO) downlink transmission.

Referring to FIG. 1B, in one embodiment of the invention, the base station 102 b may select, from a plurality of users, a first user having a channel gain that is greater than a channel gain corresponding to a remaining portion of the plurality of the users within the communication system 100 b. The base station 102 b may select at least a second user from a portion of the remaining portion of the plurality of the users. The channel direction of the selected at least the second user may be approximately orthogonal to that of the first user. The channel gain may be defined by a quantized channel gain. The base station 102 b may determine a quantized channel direction of the selected first user for the selecting the at least the second user. The base station 102 b may filter the remaining portion of the plurality of the users for the selecting the at least the second user. The filtering may be achieved by the following thresholding orthogonality function:

${flag}_{k} = \left\{ {\begin{matrix} 1 & {\alpha_{k} \geq \alpha_{th}} \\ 0 & {\alpha_{k} < \alpha_{th}} \end{matrix},{{{where}\mspace{14mu}\alpha_{k}} = \frac{\eta_{k}}{\gamma_{k}}},{\eta_{k} = {\left\langle {h_{k},{\hat{v}}_{i}^{\bot}} \right\rangle }^{2}},} \right.$ where k is an index, η_(k) is received signal to noise ratio of the k^(th) user with respect to the broadcast beam {circumflex over (v)}_(i) ^(⊥)* within the communication system 100 b, α _(th) is a threshold, α_(k) is an orthogonality parameter, and γ_(k) is a channel gain of the k^(th) user within the communication system 100 b.

The base station 102 b may be adapted to maximize a system capacity based on the selecting the first user and the selecting at least the second user. The base station 102 b may generate a first preceding matrix for precoding at least one transmit signal within the communication system, if a system capacity function for the selected first user and the selected at least the second user is greater than a system capacity function for the selected first user. The first precoding matrix may be determined from a quantized channel direction of the selected first user and an orthogonal direction of the quantized channel direction of the selected first user. The base station 102 b may generate a second preceding matrix for precoding at least one transmit signal within the communication system, if a system capacity function for the selected first user and the selected at least the second user is not greater than a system capacity function for the selected first user. The second precoding matrix is determined from a complex conjugate of a quantized channel direction of the selected first user.

Accordingly, the present invention may be realized in hardware, software, or a combination of hardware and software. The present invention may be realized in a centralized fashion in at least one computer system, or in a distributed fashion where different elements are spread across several interconnected computer systems. Any kind of computer system or other apparatus adapted for carrying out the methods described herein is suited. A typical combination of hardware and software may be a general-purpose computer system with a computer program that, when being loaded and executed, controls the computer system such that it carries out the methods described herein.

The present invention may also be embedded in a computer program product, which comprises all the features enabling the implementation of the methods described herein, and which when loaded in a computer system is able to carry out these methods. Computer program in the present context means any expression, in any language, code or notation, of a set of instructions intended to cause a system having an information processing capability to perform a particular function either directly or after either or both of the following: a) conversion to another language, code or notation; b) reproduction in a different material form.

While the present invention has been described with reference to certain embodiments, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted without departing from the scope of the present invention. In addition, many modifications may be made to adapt a particular situation or material to the teachings of the present invention without departing from its scope. Therefore, it is intended that the present invention not be limited to the particular embodiment disclosed, but that the present invention will include all embodiments falling within the scope of the appended claims. 

1. A method for processing signals in a communication system, the method comprising: selecting, from a plurality of users, a first user having a channel gain that is greater than a channel gain corresponding to a remaining portion of said plurality of users; and selecting at least a second user from a portion of said remaining portion of said plurality of users, based on a feedback channel gain of said second user, wherein said selected at least said second user has a channel direction that is approximately orthogonal to a channel direction of said first user.
 2. The method according to claim 1, wherein said channel gain is defined by a quantized channel gain.
 3. The method according to claim 1, comprising determining a quantized channel direction of said selected first user for said selecting said at least said second user.
 4. The method according to claim 1, comprising filtering said remaining portion of said plurality of users for said selecting said at least said second users.
 5. The method according to claim 4, wherein said filtering is achieved by the following thresholding orthogonality function: ${flag}_{k} = \left\{ {\begin{matrix} 1 & {\alpha_{k} \geq \alpha_{th}} \\ 0 & {\alpha_{k} < \alpha_{th}} \end{matrix},{{{where}\mspace{14mu}\alpha_{k}} = \frac{\eta_{k}}{\gamma_{k}}},{\eta_{k} = {\left\langle {h_{k},{\hat{v}i^{\bot}}} \right\rangle }^{2}},} \right.$ wherein k is an index, η_(k) is a received signal-to-noise ratio of the k^(th) user with respect to a broadcast beam {circumflex over (v)}_(i) ^(⊥)* within the communication system, α_(th) is a threshold, α_(k) is an orthogonality parameter, h_(k) is the channel vector to the k^(th) user, and γ_(k) is a channel gain of said k^(th) user within the communication system.
 6. The method according to claim 1, comprising maximizing a system capacity based on said selecting said first user and said selecting at least said second user.
 7. The method according to claim 1, comprising, if a system capacity function for said selected first user and said selected at least said second user is greater than a system capacity function for said selected first user, generating a first precoding matrix for precoding at least one transmit signal within the communication system.
 8. The method according to claim 7, wherein said first precoding matrix is determined from a quantized channel direction of said selected first user and an orthogonal direction of said quantized channel direction of said selected first user.
 9. The method according to claim 1, comprising, if a system capacity function for said selected first user and said selected at least said second user is not greater than a system capacity function for said selected first user, generating a second precoding matrix for precoding at least one transmit signal within the communication system.
 10. The method according to claim 9, wherein said second precoding matrix is determined from a complex conjugate of a quantized channel direction of said selected first user.
 11. A system for processing signals in a communication system, the system comprising: circuitry that selects, from a plurality of users, a first user having a channel gain that is greater than a channel gain corresponding to a remaining portion of said plurality of users; and said circuitry selects at least a second user from a portion of said remaining portion of said plurality of users, based on a feedback channel gain of said second user, wherein said selected at least said second user has a channel direction that is approximately orthogonal to a channel direction of said first user.
 12. The system according to claim 11, wherein said channel gain is defined by a quantized channel gain.
 13. The system according to claim 11, wherein said circuitry determines a quantized channel direction of said selected first user for said selecting said at least said second user.
 14. The system according to claim 11, wherein said circuitry filters said remaining portion of said plurality of users for said selecting said at least said second user.
 15. The system according to claim 14, wherein said filtering is achieved by the following thresholding orthogonality function: ${flag}_{k} = \left\{ {\begin{matrix} 1 & {\alpha_{k} \geq \alpha_{th}} \\ 0 & {\alpha_{k} < \alpha_{th}} \end{matrix},{{{where}\mspace{14mu}\alpha_{k}} = \frac{\eta_{k}}{\gamma_{k}}},{\eta_{k} = {\left\langle {h_{k},{\hat{v}i^{\bot}}} \right\rangle }^{2}},} \right.$ wherein k is an index, η_(k) is a received signal-to-noise ratio of the k^(th) user with respect to a broadcast beam {circumflex over (v)}_(i) ^(⊥)* within the communication system, α_(th) is a threshold, α_(k) is an orthogonality parameter, h_(k) is the channel vector to the k^(th) user, and γ_(k) is a channel gain of said k^(th) user within the communication system.
 16. The system according to claim 11, wherein said circuitry maximizes a system capacity based on said selecting said first user and said selecting at least said second user.
 17. The system according to claim 11, wherein said circuitry generates a first precoding matrix for precoding at least one transmit signal within the communication system, if a system capacity function for said selected first user and said selected at least said second user is greater than a system capacity function for said selected first user.
 18. The system according to claim 17, wherein said first precoding matrix is determined from a quantized channel direction of said selected first user and an orthogonal direction of said quantized channel direction of said selected first user.
 19. The system according to claim 1, wherein said circuitry generates a second precoding matrix for precoding at least one transmit signal within the communication system, if a system capacity function for said selected first user and said selected at least said second user is not greater than a system capacity function for said selected first user.
 20. The system according to claim 19, wherein said second precoding matrix is determined from a complex conjugate of a quantized channel direction of said selected first user.
 21. A non-transitory machine-readable storage having stored thereon, a computer program having at least one code section for processing signals in a communication system, the at least one code section being executable by a machine for causing the machine to perform steps comprising: selecting, from a plurality of users, a first user having a channel gain that is greater than a channel gain corresponding to a remaining portion of said plurality of users; and selecting at least a second user from a portion of said remaining portion of said plurality of users, based on a feedback channel gain of said second user, wherein said selected at least said second user has a channel direction that is approximately orthogonal to a channel direction of said first user.
 22. The machine-readable storage according to claim 21, wherein said channel gain is defined by a quantized channel gain.
 23. The machine-readable storage according to claim 21, comprising code for determining a quantized channel direction of said selected first user for said selecting said at least said second user.
 24. The machine-readable storage according to claim 21, comprising code for filtering said remaining portion of said plurality of users for said selecting said at least said second user.
 25. The machine-readable storage according to claim 24, wherein said filtering is achieved by the following thresholding orthogonality function: ${flag}_{k} = \left\{ {\begin{matrix} 1 & {\alpha_{k} \geq \alpha_{th}} \\ 0 & {\alpha_{k} < \alpha_{th}} \end{matrix},{{{where}\mspace{14mu}\alpha_{k}} = \frac{\eta_{k}}{\gamma_{k}}},{\eta_{k} = {\left\langle {h_{k},{\hat{v}i^{\bot}}} \right\rangle }^{2}},} \right.$ wherein k is an index, η_(k) is a received signal-to-noise ratio of the k^(th) user with respect to a broadcast beam {circumflex over (v)}_(i) ^(⊥)* within the communication system, α_(th) is a threshold, α_(k) is an orthogonality parameter, h_(k) is the channel vector to the k^(th) user, and γ_(k) is a channel gain of said k^(th) user within the communication system.
 26. The machine-readable storage according to claim 21, comprising code for maximizing a system capacity based on said selecting said first user and said selecting at least said second user.
 27. The machine-readable storage according to claim 21, comprising code for generating a first precoding matrix for precoding at least one transmit signal within the communication system, if a system capacity function for said selected first user and said selected at least said second user is greater than a system capacity function for said selected first user.
 28. The machine-readable storage according to claim 27, wherein said first precoding matrix is determined from a quantized channel direction of said selected first user and a quantized orthogonal direction of said selected first user.
 29. The machine-readable storage according to claim 21, comprising code for generating a second precoding matrix for precoding at least one transmit signal within the communication system, if a system capacity function for said selected first user and said selected at least said second user is not greater than a system capacity function for said selected first user.
 30. The machine-readable storage according to claim 29, wherein said second precoding matrix is determined from a complex conjugate of a quantized channel direction of said selected first user. 